Radar device

ABSTRACT

A radar transmitter transmits a radar signal through a transmitting array antenna at a predetermined transmission period, and a radar receiver receives a reflected wave signal which is the radar signal reflected by a target through a receiving array antenna. A transmitting array antenna and a receiving array antenna each include multiple subarray elements, the subarray elements in the transmitting array antenna and the receiving array antenna are linearly arranged in a first direction, each subarray element includes multiple antenna elements, the subarray element has a dimension larger than a predetermined antenna element spacing in the first direction, and an absolute value of a difference between a subarray element spacing of the transmitting array antenna and a subarray element spacing of the receiving array antenna is equal to the predetermined antenna element spacing.

BACKGROUND

1. Technical Field

The present disclosure relates to a radar device.

2. Description of the Related Art

Recent studies have been made on a radar device using a radartransmission signal of a short wavelength including microwave ormilliwave with which a high resolution is achieved. Development isrequired on a radar device (wide-angle radar device) that detectsobjects (targets) including a pedestrian as well as a vehicle in awide-angle range in order to improve outdoor safety.

A known example of such a radar device is a pulse radar device thatrepeatedly emits pulsed waves. A wide-angle pulse radar that detects avehicle and a pedestrian in a wide-angle range receives a mixture ofmultiple reflected waves from a target (for example, a vehicle) at ashort distance and a target (for example, a pedestrian) at a longdistance. This requires (1) a radar transmitter to have a configurationto transmit pulsed waves or pulse-modulated waves having aauto-correlation characteristic (hereinafter, referred to as a low rangesidelobe characteristic) that achieves low range sidelobes, and (2) aradar receiver to have a configuration with a wide reception dynamicrange.

Examples of the configuration of the wide-angle radar device include thefollowing two configurations.

The first configuration transmits pulsed waves or modulated waves asradar waves by mechanical or electrical scanning using a directionalbeam of a narrow angle (beam width of a few degrees), and receivesreflected waves using a narrow-angle directional beam. With thisconfiguration, the scanning needs to be performed a large number oftimes to obtain a high resolution, which leads to a degradation in theperformance of following a fast moving target.

The second configuration uses a method (direction of arrival (DOA)estimation) of receiving reflected waves through an array antennaincluding multiple antennas (antenna elements), and estimating thearrival angle of the reflected waves using a signal processing algorithmbased on a reception phase difference corresponding to an antennaspacing. This configuration allows the radar receiver to estimate thearrival angle even when a frequency of scanning of a transmission beamon the radar transmitter is reduced, thereby achieving a shortenedscanning time and an improved following performance as compared to thefirst configuration. Examples of DOA estimation methods include aFourier transform based on matrix calculation, a Capon method and alinear prediction (LP) method based on inverse matrix calculation, and amultiple signal classification (MUSIC) and an estimation of signalparameters via rotational invariance techniques (ESPRIT) based oneigenvalue calculation.

Disclosed is a radar device (also referred to as a MIMO radar) thatincludes multiple antennas (array antennas) on the radar transmitter aswell as the radar receiver and performs beam scanning by signalprocessing using transmitting and receiving array antennas (see Jian Li,Stoica, Petre, “MIMO Radar with Colocated Antennas,” Signal ProcessingMagazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007, for example).

In order to achieve a high directional gain of an array antenna, antennaelements (hereinafter, referred to as array elements) included in thearray antenna are each formed of a subarray antenna including multipleantenna elements in some cases.

As for the element spacing of the array antenna, it is difficult toarrange the array elements at spacings smaller than the size of thearray element. However, the dimension of the array element having asubarray antenna configuration is large, and accordingly a large spacingis needed between subarray antennas, which may generate a grating lobeon a directivity pattern of the array antenna.

SUMMARY

One non-limiting and exemplary embodiment of the present disclosureprovides a radar device that can prevent generation of an unnecessarygrating lobe and achieve a desired directivity pattern even whenarranged in the subarray antenna configuration.

In one general aspect of the present disclosure, the techniquesdisclosed here feature a radar device including: radar transmissioncircuitry which, in operation, transmits a radar signal through atransmitting array antenna at a predetermined transmission period; andradar reception circuitry which, in operation, receives a reflected wavesignal which is the radar signal reflected by an object through areceiving array antenna. The transmitting array antenna and thereceiving array antenna each include multiple subarray elements. Thesubarray elements are linearly arranged in a first direction in each ofthe transmitting array antenna and the receiving array antenna. Eachsubarray element includes multiple antenna elements. A dimension of eachsubarray element in the first direction is larger than a predeterminedantenna element spacing. An absolute value of a difference between asubarray element spacing of the transmitting array antenna and asubarray element spacing of the receiving array antenna is equal to thepredetermined antenna element spacing.

One aspect of the present disclosure can prevent generation of anunnecessary grating lobe and achieve a desired directivity pattern evenin a subarray antenna configuration.

It should be noted that general or specific embodiments may beimplemented as a system, a method, an integrated circuit, a computerprogram, a storage medium, or any selective combination thereof.

Additional benefits and advantages of the disclosed embodiments willbecome apparent from the specification and drawings. The benefits and/oradvantages may be individually obtained by the various embodiments andfeatures of the specification and drawings, which need not all beprovided in order to obtain one or more of such benefits and/oradvantages.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A illustrates an exemplary configuration of a subarray element;

FIG. 1B illustrates an exemplary configuration of an array antennaincluding the subarray elements;

FIG. 2 is a block diagram of the configuration of a radar deviceaccording to an embodiment of the present disclosure;

FIG. 3 illustrates an exemplary radar transmission signal according tothe embodiment of the present disclosure;

FIG. 4 is a block diagram of another configuration of a radartransmission signal generator according to the embodiment of the presentdisclosure;

FIG. 5 illustrates an example of a transmission timing and a measurementrange of the radar transmission signal according to the embodiment ofthe present disclosure;

FIG. 6 illustrates antenna arrangements of a transmitting array, areceiving array, and a virtual receiving array according to theembodiment of the present disclosure;

FIG. 7 illustrates a directivity pattern according to the embodiment ofthe present disclosure;

FIG. 8 illustrates antenna arrangements of a transmitting array, areceiving array, and a virtual receiving array according to Variation 1of the embodiment of the present disclosure;

FIG. 9A illustrates a directivity pattern in the horizontal directionaccording to Variation 1 of the embodiment of the present disclosure;

FIG. 9B illustrates a directivity pattern in the vertical directionaccording to Variation 1 of the embodiment of the present disclosure;

FIG. 10 illustrates antenna arrangements of a transmitting array, areceiving array, and a virtual receiving array according to Variation 2of the embodiment of the present disclosure;

FIG. 11A illustrates a directivity pattern in the horizontal directionaccording to Variation 2 of the embodiment of the present disclosure;

FIG. 11B illustrates a directivity pattern in the vertical directionaccording to Variation 2 of the embodiment of the present disclosure;

FIG. 12 illustrates antenna arrangements of a transmitting array, areceiving array, and a virtual receiving array according to Variation 3of the embodiment of the present disclosure;

FIG. 13A illustrates a directivity pattern in the horizontal directionaccording to Variation 3 of the embodiment of the present disclosure;and

FIG. 13B illustrates a directivity pattern in the vertical directionaccording to Variation 3 of the embodiment of the present disclosure.

DETAILED DESCRIPTION [Underlying Knowledge Forming Basis of the PresentDisclosure]

FIG. 1A illustrates an exemplary subarray configuration (hereinafter,also referred to as a subarray element) including an antenna element.The subarray element illustrated in FIG. 1A includes four antennaelements in a 2×2 matrix. In the example illustrated in FIG. 1A, thesubarray element has a dimension of 0.8 wavelength in both thehorizontal and vertical directions.

FIG. 1B illustrates an exemplary array antenna including a lineararrangement of the four subarray elements illustrated in FIG. 1A. Sinceeach subarray element has a dimension of 0.8 wavelength (refer to FIG.1A) as illustrated in FIG. 1B, a spacing between the subarray elementsneeds to be about 1 wavelength or larger.

For example, an array element spacing (predetermined element spacing)for preventing generation of a grating lobe in a range of ±90° of a mainlobe is 0.5 wavelength. In the array antenna illustrated in FIG. 1B, thesubarray elements have an element spacing of about 1 wavelength orlarger, and thus it is difficult to obtain a predetermined elementspacing. This results in generation of a grating lobe in the range of±90° of the main lobe.

As described above, when the dimension of the subarray element is 0.5wavelength or larger, it may be difficult to set the element spacing ofthe array antenna to be 0.5 wavelength. Thus, an unnecessary gratinglobe occurs in the range of ±90° of the main lobe, which generates avirtual image at angle determination, thereby leading to a falsedetection.

Published Japanese Translation of PCT International Application No.2011-526370 (hereinafter, referred to as Patent Document 1) discloses anarray antenna configuration including a subarray element having a widthd equal to 1 wavelength approximately. In Patent Document 1,transmitting antennas Tx0 and Tx1 has an element spacing of 6wavelengths, and receiving antennas RX0, RX1, RX2, and RX3 has anelement spacing of 1.5 wavelengths ±(λ/8) (where λ represents 1wavelength). With the array antenna configuration disclosed in PatentDocument 1, time-divisional transmission of a radar transmission signalis performed through the transmitting antennas Tx0 and Tx1, and areception signal for the radar transmission signal transmitted from eachof the transmitting antennas Tx0 and Tx1 is acquired through thereceiving antennas RX0, RX1, RX2, and RX3.

With this configuration, a phase difference due to the difference of thepositions of the transmitting antennas is superimposed on the receptionsignal acquired by a receiving array antenna. This achieves a virtuallyincreased opening length of the receiving antennas. Hereinafter, avirtual receiving array antenna having an increased effective openinglength due to antenna element arrangements of transmitting and receivingarray antennas is referred to as a “virtual receiving array”.

In Patent Document 1, however, the receiving array antennas have anelement spacing of 1.5 wavelengths ±λ/8, which leads to generation of agrating lobe in a direction shifted from a main beam direction by 40°approximately.

An aspect according to the present disclosure prevents generation of anunnecessary grating lobe and achieves a desired directivity pattern evenwhen array elements having a subarray configuration are arranged.

Hereinafter, embodiments according to the aspect of the presentdisclosure are described in detail with reference to the accompanyingdrawings. An identical reference numeral is given to any identicalcomponent common to the embodiments, and description thereof is omittedto avoid duplication.

[Radar Device Configuration]

FIG. 2 is a block diagram of the configuration of a radar device 10according to the present embodiment.

The radar device 10 includes a radar transmitter 100, a radar receiver200, and a reference signal generator 300.

The radar transmitter 100 generates a radar signal (radar transmissionsignal) of a high frequency based on a reference signal received fromthe reference signal generator 300. Then, the radar transmitter 100transmits the radar transmission signal at a predetermined transmissionperiod through a transmitting array antenna that includes multipletransmitting antennas 106-1 to 106-Nt.

The radar receiver 200 receives a reflected wave signal which is theradar transmission signal reflected by a target (not illustrated)through a receiving array antenna that includes multiple receivingantennas 202-1 to 202-Na. The radar receiver 200 processes the reflectedwave signal received through the antennas 202 based on the referencesignal received from the reference signal generator 300 so as toperform, for example, an existence detection and a direction estimationof the target. The target is an object to be detected by the radardevice 10, and examples thereof include a vehicle and a person.

The reference signal generator 300 is connected to the radar transmitter100 and the radar receiver 200. The reference signal generator 300supplies the common reference signal to both of the radar transmitter100 and the radar receiver 200 to synchronize processing of the radartransmitter 100 and the radar receiver 200.

[Configuration of Radar Transmitter 100]

The radar transmitter 100 includes radar transmission signal generators101-1 to 101-Nt, transmission radio units 105-1 to 105-Nt, and thetransmitting antennas 106-1 to 106-Nt. Thus, the radar transmitter 100includes Nt transmitting antennas 106, and each transmitting antenna 106is connected to the corresponding radar transmission signal generator101 and the corresponding transmission radio unit 105.

The radar transmission signal generator 101 generates a timing clockobtained by multiplying the reference signal received from the referencesignal generator 300 by a predetermined value, and generates the radartransmission signal based on the generated timing clock. Then, the radartransmission signal generator 101 repeatedly outputs the radartransmission signal at a predetermined radar transmission period (Tr).The radar transmission signal is expressed in r_(z)(k, M)=I_(Z)(k,M)+jQ_(Z)(k, M). In this expression, z represents an index correspondingto each transmitting antenna 106, and is z=1, . . . , Nt, and jrepresents the imaginary unit, k represents a discrete time, and Mrepresents an ordinal number of the radar transmission period.

Each radar transmission signal generator 101 includes a code generator102, a modulator 103, and a low pass filter (LPF) 104. The followingdescribes components of the radar transmission signal generator 101-zcorresponding to the z-th (z=1, . . . , Nt) transmitting antenna 106.

Specifically, the code generator 102 generates codes a(z)_(n) (n=1, . .. , L) (pulse codes) in a code sequence of a code length L at each radartransmission period Tr. Codes having low correlation or no correlationtherebetween are used as the codes a(z)_(n) (z=1, . . . , Nt) generatedby each of the code generators 102-1 to 102-Nt. Examples of the codesequence include a Walsh-Hadamard code, an M-sequence code, and a Goldcode.

The modulator 103 provides the codes a(z)_(n) received from the codegenerator 102 with a pulse modulation (amplitude modulation, amplitudeshift keying (ASK), and pulse shift keying) or a phase modulation (phaseshift keying), and outputs a modulated signal to the LPF 104.

The LPF 104 outputs a signal component of the modulated signal receivedfrom the modulator 103, which is not higher than a predeterminedthreshold band, as a baseband radar transmission signal to thetransmission radio unit 105.

The z-th (z=1, . . . , Nt) transmission radio unit 105 performs afrequency conversion on the baseband radar transmission signal outputfrom the z-th radar transmission signal generator 101, generates a radartransmission signal in a carrier frequency (radio frequency or RF) band,amplifies this radar transmission signal through a transmissionamplifier to have a predetermined transmission electric power P [dB],and outputs the amplified radar transmission signal to the z-thtransmitting antenna 106.

The z-th (z=1, . . . , Nt) transmitting antenna 106 emits the radartransmission signal output from the z-th transmission radio unit 105into space.

FIG. 3 illustrates the radar transmission signal transmitted from the Nttransmitting antennas 106 of the radar transmitter 100. A codetransmission slot Tw includes a pulse code sequence of the code lengthL. In each radar transmission period Tr, the pulse code sequence istransmitted in the code transmission slot Tw, and no signal istransmitted in the remaining slot (Tr−Tw). A pulse modulation using Nosamples is performed per one pulse code (a(z)_(n)), and thus Nr (=No×L)sample signals are included in each code transmission slot Tw. In otherwords, the modulator 103 employs a sampling rate of (No×L)/Tw. The slot(Tr−Tw) with no signal includes Nu samples.

The radar transmitter 100 includes, in place of the radar transmissionsignal generator 101, a radar transmission signal generator 101 aillustrated in FIG. 4. The radar transmission signal generator 101 adoes not include the code generator 102, the modulator 103, or the LPF104, which are illustrated in FIG. 2, but includes a code storage 111and a DA converter 112 instead. The code storage 111 previously stores acode sequence generated by the code generator 102 (FIG. 2), andcyclically and sequentially reads out the stored code sequence. The DAconverter 112 converts a code sequence (digital signal) output from thecode storage 111 into an analog signal.

[Configuration of Radar Receiver 200]

In FIG. 2, the radar receiver 200 includes Na receiving antennas 202,constituting an array antenna. The radar receiver 200 also includes Naantenna system processors 201-1 to 201-Na and a direction estimator 214.

Each receiving antenna 202 receives the reflected wave signal which isthe radar transmission signal reflected by the target (object), andoutputs the reflected wave signal thus received as a reception signal tothe corresponding antenna system processor 201.

Each antenna system processor 201 includes a reception radio unit 203and a signal processor 207.

The reception radio unit 203 includes an amplifier 204, a frequencyconverter 205, and an quadrature detector 206. The reception radio unit203 generates a timing clock obtained by multiplying the referencesignal received from the reference signal generator 300 by apredetermined value, and operates based on this generated timing clock.Specifically, the amplifier 204 amplifies the reception signal receivedfrom the receiving antenna 202 to a predetermined level, and thefrequency converter 205 performs a frequency conversion of a highfrequency band of the reception signal into a baseband. Then, thequadrature detector 206 converts this reception signal in the basebandinto a reception signal in a baseband including an I signal and a Qsignal.

The signal processor 207 includes AD converters 208 and 209 andseparators 210-1 to 210-Nt.

The AD converter 208 receives the I signal from the quadrature detector206, and the AD converter 209 receives the Q signal from the quadraturedetector 206. The AD converter 208 performs discrete time sampling on abaseband signal including the I signal so as to convert the I signalinto digital data. The AD converter 209 performs discrete time samplingon a baseband signal including the Q signal so as to convert the Qsignal into digital data.

The samplings performed by the AD converters 208 and 209 include Nsdiscrete samplings in a duration Tp (=Tw/L) of one sub pulse in theradar transmission signal. In other words, Ns oversamplings areperformed per one sub pulse.

The following description uses an I signal I_(r)(k, M) and a Q signalQ_(r)(k, M) to express a baseband reception signal as the output fromthe AD converters 208 and 209 at the discrete time k in the M-th radartransmission period Tr[M], as a complex signal x(k, M)=I_(r)(k,M)+jQ_(r)(k, M). In the following, the discrete time k is defined withreference to a timing (k=1) at which the radar transmission period (Tr)starts, and the signal processor 207 periodically operates untilk=(Nr+Nu)Ns/No, which is a last sample point before the radartransmission period Tr ends. Namely, the discrete time k has a value ofk=1, . . . , (Nr+Nu)Ns/No, where j is the imaginary unit.

The signal processor 207 includes the Nt separators 210 whose number isequal to the number of systems as the number of the transmittingantennas 106. Each separator 210 includes a correlation calculator 211,an adder 212, and a Doppler frequency analyzer 213. The configuration ofthe z-th (z=1, . . . , Nt) separator 210 is described below.

The correlation calculator 211 calculates a correlation between thediscrete sample value x(k, M) and the pulse codes a(z)_(n) (where z=1, .. . , Nt, and n=1, . . . , L) of the code length L transmitted throughthe radar transmitter 100. The discrete sample value x(k, M) includesthe discrete sample values I_(r)(k, M) and Q_(r)(k, M) received from theAD converters 208 and 209 at each radar transmission period Tr. Forexample, the correlation calculator 211 calculates a sliding correlationbetween the discrete sample value x(k, M) and the pulse codes a(z)_(n).For example, a correlation calculated value AC_((z))(k, M) for thesliding correlation at the discrete time k in the M-th radartransmission period Tr[M] is calculated based on an expression below.

$\begin{matrix}{{A\; {C_{(z)}\left( {k,M} \right)}} = {\sum\limits_{n = 1}^{L}{{x\left( {{k + {N_{s}\left( {n - 1} \right)}},M} \right)}{a(z)}_{n}^{*}}}} & (1)\end{matrix}$

In the above expression, an asterisk (*) represents a complex conjugateoperator.

The correlation calculator 211 calculates the correlation over aduration of k=1, . . . , (Nr+Nu)Ns/No in accordance with, for example,Expression (1).

The correlation calculator 211 is not limited to the correlationcalculation for k=1, . . . , (Nr+Nu)Ns/No, and may limit a measurementrange (that is, the range of k) depending on the range of existence ofthe target as a measurement target of the radar device 10. This allowsthe radar device 10 to achieve a reduction in the calculation amount ofthe correlation calculator 211. For example, the correlation calculator211 may limit the measurement range to k=Ns(L+1), . . . ,(Nr+Nu)Ns/No−NsL. In this case, as illustrated in FIG. 5, the radardevice 10 does not perform measurement in the time slot corresponding tothe code transmission slot Tw.

Thus, in the radar device 10, the correlation calculator 211 does notperform processing in a duration (at least a duration less than τ1) inwhich the radar transmission signal going directly into the radarreceiver 200, which achieves a measurement free from the influence ofthis phenomenon. When the measurement range (range of k) of thecorrelation calculation is limited, the measurement range (range of k)of processing at the adder 212, the Doppler frequency analyzer 213, andthe direction estimator 214 to be described later may be limited aswell. This achieves a reduced processing amount at each component andthus a reduced electric power consumption at the radar receiver 200.

At each discrete time k in the M-th radar transmission period Tr, theadder 212 performs an addition (coherent integration) of the correlationcalculated value AC_((z))(k, M) received from the correlation calculator211 over a duration (Tr×Np) that is a predetermined number (Np) of theradar transmission periods Tr. This addition (coherent integration)processing involving the number Np of additions over the duration(Tr×Np) is performed by an expression below.

$\begin{matrix}{{{CI}_{(z)}\left( {k,m} \right)} = {\sum\limits_{g = 1}^{N_{p}}{A\; {C_{(z)}\left( {k,{{N_{p}\left( {m - 1} \right)} + g}} \right)}}}} & (2)\end{matrix}$

In this expression, Cl_((z))(k, m) represents the sum (hereinafter, alsoreferred to as the correlation sum) of the correlation calculatedvalues, where Np is an integer equal to or larger than one, and m is aninteger equal to or larger than one as an ordinal number of the numberof additions in units of the number Np of additions at the adder 212.The value z is z=1, . . . , Nt.

The adder 212 performs Np additions in units of the output from thecorrelation calculator 211 obtained per the radar transmission periodTr. Specifically, the adder 212 calculates the correlation valueCl_((z))(k, m) as the sum of the correlation calculated valuesAC_((z))(k, Np(m−1)+1) to AC_((z))(k, Np×m) for each discrete time k.Thus, the adder 212 can achieve an improved SNR of the reflected wavesignal through the Np additions of the correlation calculated value in arange in which the reflected wave signals from the target have a highcorrelation. This then achieves an improved measurement performancerelated to estimation of an arrival distance of the target.

In order to obtain an ideal addition gain, a condition for having aphase component of each correlation calculated value within a certainrange needs to be satisfied in an addition time period for the number Npof additions of the correlation calculated value. Thus, the number Np ofadditions is preferably set based on an expected maximum moving speed ofthe target as a measurement target. This is because a larger expectedmaximum speed of the target results in a larger amount of change of aDoppler frequency included in a reflected wave from the target, and thusa shorter time duration in which a high correlation is obtained. In thiscase, the number Np of additions is small, which leads to a small gainimprovement effect of the addition by the adder 212.

The Doppler frequency analyzer 213 performs the coherent integration forthe same discrete time k in units of Cl_((z))(k, Nc(w−1)+1) toCl_((z))(k, Nc×w) as Nc outputs obtained at each discrete time k fromthe adder 212. For example, the Doppler frequency analyzer 213 correctsa phase variation φ(fs)=2πfs(Tr×Np)Δφ in accordance with 2Nf differentDoppler frequencies fsΔφ as expressed in an expression below, and thenperforms the coherent integration.

$\begin{matrix}\begin{matrix}{{{FT\_ CI}_{(z)}^{Nant}\left( {k,f_{s},w} \right)} = {\sum\limits_{q = 0}^{N_{c} - 1}{{CI}_{(z)}\left( {k,{{N_{c}\left( {w - 1} \right)} + q + 1}} \right)}}} \\{{\exp \left\lbrack {{- j}\; {\varphi \left( f_{s} \right)}q} \right\rbrack}} \\{= {\sum\limits_{q = 0}^{N_{c} - 1}{{CI}_{(z)}\left( {k,{{N_{c}\left( {w - 1} \right)} + q + 1}} \right)}}} \\{{\exp \left\lbrack {{- j}\; 2\; \pi \; f_{s}T_{r}N_{p}q\; \Delta \; \varphi} \right\rbrack}}\end{matrix} & (3)\end{matrix}$

In this expression, FT_Cl_((z)) ^(Nant)(k, fs, w) represents the w-thoutput of the Doppler frequency analyzer 213, and represents a result ofthe coherent integration of the Doppler frequency fsΔφ by the Nant-thantenna system processor 201 at the discrete time k. Also in theexpression, Nant=1 to Na, fs=−Nf+1, . . . , 0, . . . , Nf, k=1, . . . ,(Nr+Nu)Ns/No, w is an integer equal to or larger than one, and Δφrepresents a phase rotation unit.

Accorindgly, each antenna system processor 201 obtains FT_Cl_((z))^(Nant)(k, −Nf+1, w), . . . , FT_Cl_((z)) ^(Nant)(k, Nf−1, w) as theresult of the coherent integration in accordance with 2Nf Dopplerfrequency components at the discrete time k, for each duration(Tr×Np×Nc) of a plurality, Np×Nc, of the radar transmission periods Tr.In the expression, j is the imaginary unit, and the value z is z=1, . .. , Nt.

For Δφ=1/Nc, the above-described processing of the Doppler frequencyanalyzer 213 is equivalent to discrete Fourier transform (DFT)processing of the outputs from the adder 212 with a sampling intervalTm=(Tr×Np) and a sampling frequency fm=1/Tm.

Setting Nf to a power of two allows application of fast Fouriertransform (FFT) processing at the Doppler frequency analyzer 213,thereby achieving a large reduction in the calculation amount. ForNf>Nc, the FFT processing can also be applied in a region in which q>Ncholds by performing zero filling processing that sets Cl_((z))(k,Nc(w−1)+q)=0, and thus the calculation amount can be largely reducedaccordingly.

The Doppler frequency analyzer 213 may perform, in place of the FFTprocessing, processing of sequentially calculating the product sumcalculation expressed in Expression (3) above. Specifically, the Dopplerfrequency analyzer 213 may generate a coefficient exp[−j2πf_(s)T_(r)N_(p)qΔφ] for fs=−Nf+1, . . . , 0, . . . , Nf−1 forCl_((z))(k, Nc(w−1)+q+1) as the Nc outputs obtained from the adder 212at each discrete time k, and sequentially perform the product sumcalculation processing. Here, q=0 to Nc−1.

In the following, a virtual receiving array correlation vector h(k, fs,w) in expressions below represents a set of the w-th outputs FT_Cl_((z))¹(k, fs, w), FT_Cl_((z)) ²(k, fs, w), . . . , FT_Cl_((z)) ^(Na)(k, fs,w) obtained through the same processing on the Na respective antennasystem processors 201. The virtual receiving array correlation vectorh(k, fs, w) includes Nt×Na elements as the product of the number oftransmitting antennas, Nt, and the number of receiving antennas, Na. Thevirtual receiving array correlation vector h(k, fs, w) is used to laterdescribe the processing of performing a direction estimation of thereflected wave signal from the target based on a phase differencebetween the receiving antennas 202. Here, z=1, . . . , Nt, and b=1, . .. , Na.

$\begin{matrix}{{h\left( {k,{fs},w} \right)} = {\begin{bmatrix}{{FT\_ CI}_{(1)}^{1}\left( {k,{fs},w} \right)} \\{{FT\_ CI}_{(2)}^{1}\left( {k,{fs},w} \right)} \\\vdots \\{{FT\_ CI}_{({Nt})}^{1}\left( {k,{fs},w} \right)} \\{{FT\_ CI}_{(1)}^{2}\left( {k,{fs},w} \right)} \\{{FT\_ CI}_{(2)}^{2}\left( {k,{fs},w} \right)} \\\vdots \\{{FT\_ CI}_{({Nt})}^{2}\left( {k,{fs},w} \right)} \\\vdots \\{{FT\_ CI}_{(1)}^{Na}\left( {k,{fs},w} \right)} \\{{FT\_ CI}_{(2)}^{Na}\left( {k,{fs},w} \right)} \\\vdots \\{{FT\_ CI}_{({Nt})}^{Na}\left( {k,{fs},w} \right)}\end{bmatrix} = \begin{bmatrix}{h^{1}\left( {k,{fs},w} \right)} \\{h^{2}\left( {k,{fs},w} \right)} \\\vdots \\{h^{Na}\left( {k,{fs},w} \right)}\end{bmatrix}}} & (4) \\{{h^{b}\left( {k,{fs},w} \right)} = \begin{bmatrix}{{FT\_ CI}_{(1)}^{b}\left( {k,{fs},w} \right)} \\{{FT\_ CI}_{(2)}^{b}\left( {k,{fs},w} \right)} \\\vdots \\{{FT\_ CI}_{({Nt})}^{b}\left( {k,{fs},w} \right)}\end{bmatrix}} & (5)\end{matrix}$

In the above, the processing at each component of the signal processor207 has been described.

The direction estimator 214 calculates, for the virtual receiving arraycorrelation vector h(k, fs, w) of the w-th Doppler frequency analyzer213 output from the antenna system processors 201-1 to 201-Na, a virtualreceiving array correlation vector h_(—after—cal)(k, fs, w) obtained bycorrecting a phase deviation and an amplitude deviation between theantenna system processors 201 using an array correction valueh_cal_([y]). The virtual receiving array correlation vectorh_(—after—cal)(k, fs, w) is given by an expression below. Here, y=1, . .. , (Nt×Na).

$\begin{matrix}{{h_{{\_ after}{\_ cal}}\left( {k,{fs},w} \right)} = {{CA}\; {h\left( {k,{fs},w} \right)}}} & (6) \\{{CA} = \begin{bmatrix}{h\_ cal}_{\lbrack 1\rbrack} & 0 & \ldots & 0 \\0 & {h\_ cal}_{\lbrack 2\rbrack} & \ddots & \ldots \\\vdots & \ddots & \ddots & 0 \\0 & \ldots & 0 & {h\_ cal}_{\lbrack{{Nt} \times {Na}}\rbrack}\end{bmatrix}} & \;\end{matrix}$

Then, using the virtual receiving array correlation vectorh_(—after—cal)(k, fs, w), the direction estimator 214 performs directionestimation processing of the horizontal direction and the verticaldirection based on a phase difference of the reflected wave signalsbetween the receiving antennas 202. The direction estimator 214calculates a space profile by treating an azimuth direction θ and anelevation direction φ in a direction estimation evaluating functionvalue P (θ, φ, k, fs, w) as variables in a predetermined angle range,extracts a predetermined number of local maximum peaks of the calculatedspace profile in descending order, and sets the azimuth and elevationdirections of each local maximum peak as arrival direction estimationvalues.

Different kinds of the evaluating function value P(θ, φ, k, fs, w) areprovided by different arrival direction estimation algorithms. Forexample, an estimation method using an array antenna disclosed inCadzow, J. A., “Direction-of-arrival estimation using signal subspacemodeling,” Aerospace and Electronic Systems, IEEE Transactions onVolume: 28, Issue: 1, pp. 64-79, Publication Year: 1992.

For example, a beamformer method can be expressed in an expressionbelow. Alternatively, the Capon and MUSIC methods are applicable aswell.

P(θ_(u),φ_(v)k,fs,w)

a(θ_(u),φ_(v))^(H)h_(—after—cal)(k,fs,w)|²   (7)

In this expression, a superscript H is the Hermite transpositionoperator, and a(θ_(u), φ_(v)) represents the directional vector of thevirtual receiving array for arrival wave in an azimuth direction θ_(u)and an elevation direction φ_(v).

As described above, the direction estimator 214 outputs the w-thcalculated arrival direction estimation value, the discrete time k, theDoppler frequency fsΔφ, and the angle θ_(u), as a radar positioningresult.

The directional vector a(θ_(u), φ_(v)) is a (Nt×Na) column vectorincluding an element as a complex response of the virtual receivingarray when a reflected wave of the radar transmission signal arrives inthe azimuth direction θ_(u) and the elevation direction φ_(v). Thecomplex response a(θ_(u), φ_(v)) of the virtual receiving arrayrepresents a phase difference geometric-optically calculated dependingon an element spacing between antennas.

θ_(u) takes values separated by a predetermined azimuth spacing β₁ in anazimuth range in which the arrival direction estimation is to beperformed. For example, θ_(u) is set as described below.

θ_(u) 32 θmin+uβ ₁where u=0, . . . , NU

NU=floor[θmax−θmin)/β₁]+1

In this expression, floor(x) is a function that returns a largestinteger value not larger than a real number x.

φ_(v) takes values separated by a predetermined elevation spacing β₂ inan elevation angle range in which the arrival direction estimation is tobe performed. For example, φ_(v) is set as described below.

φ_(v)=φmin+vβ ₂ where v=0, . . . , NV

NV=floor[(φmax−φmin)/β₂]+1

The present embodiment assumes that the directional vector of thevirtual receiving array is previously calculated based on a virtualreceiving array arrangement VA#1, . . . , VA#(Nt×Na) described later.Each element of the directional vector of the virtual receiving arrayindicates a phase difference geometric-optically calculated at theelement spacing between antennas in the order of the virtual receivingarray arrangement VA#1, . . . , VA#(Nt×Na) to be described later.

The time information k described above may be converted into distanceinformation for outputting. An expression below may be used to convertthe time information k into distance information R(k). Here, Twrepresents the code transmission slot, L represents the pulse codelength, and C₀ represents the speed of light.

$\begin{matrix}{{R(k)} = {k\frac{T_{w}C_{0}}{2\; L}}} & (8)\end{matrix}$

The Doppler frequency information (fsΔφ) may be converted into arelative speed component for outputting. An expression below may be usedto convert the Doppler frequency fsΔφ into a relative speed componentvd(fs). Here, λ represents the wavelength of the RF signal output fromthe transmission radio unit 107 at a carrier frequency.

$\begin{matrix}{{v_{d}\left( f_{s} \right)} = {\frac{\lambda}{2}f_{s}\Delta \; \theta}} & (9)\end{matrix}$

[Antenna Arrangement in Radar Device 10]

Description will be made of an arrangement of the Nt transmittingantennas 106 and the Na receiving antennas 202 in the radar device 10having the configuration described above.

FIG. 6 illustrates the antenna arrangement of a transmitting arrayincluding Nt=2 transmitting antennas 106 (Tx#1 and Tx#2), the antennaarrangement of a receiving array including Na=3 receiving antennas 202(Rx#1, Rx#2, and Rx#3), and the antenna arrangement of a virtualreceiving array (including Nt x Na=6 elements) configured in accordancewith these transmitting and receiving array antennas.

Each of the transmitting antennas 106 and the receiving antennas 202 isa subarray element including two antenna elements.

D_(subarray) represents the dimension (width) of the subarray element,and De represents a predetermined antenna element spacing with which nograting lobe is generated in a radar detection angle range. In FIG. 6,the dimension D_(subarray) of the subarray element is larger than thepredetermined antenna element spacing De (D_(subarray)>De). Thepredetermined antenna element spacing De is set to be equal to or largerthan 0.5 wavelength and equal to or smaller than 0.75 wavelength.

Dt represents the subarray element spacing of the transmitting arrayantenna, and Dr represents the subarray element spacing of the receivingarray antenna. For example, in FIG. 6, the subarray element spacing Dtof the transmitting array antenna is 1.5λ (1.5 wavelength), and thesubarray element spacing Dr of the receiving array antenna is 1λ (1wavelength). Thus, the subarray element spacings Dt and Dr are equal toor larger than 1 wavelength (λ) approximately.

In the present embodiment, the dimension D_(subarray) of the subarrayelement is larger than the predetermined antenna element spacing De withwhich no grating lobe is generated in the radar detection angle range(D_(subarray)>De). In this case, the transmitting array and thereceiving array are arranged such that the subarray element spacing Dtof the transmitting array antenna and the subarray element spacing Dr ofthe receiving array antenna satisfy a relation expressed in anexpression below.

|Dt−Dr|=De   (10)

In other words, the absolute value of the difference between thesubarray element spacing Dt of the transmitting array antenna and thesubarray element spacing Dr of the receiving array antenna is equal tothe predetermined antenna element spacing De.

FIG. 6 illustrates an example case in which De is λ/2, the subarrayelement spacing Dt of the transmitting array antenna is 1.5λ, and thesubarray element spacing Dr of the receiving array antenna is λ.

In this case, as illustrated in FIG. 6, the element spacing in a centralpart (other than edges) of the virtual receiving array is equal to thepredetermined antenna element spacing De (=|Dt−Dr|=λ/2). In other words,the virtual receiving array has an array arrangement with which nograting lobe is generated in the radar detection angle range.

FIG. 7 illustrates a directivity pattern (Fourier beam pattern with themain beam at the direction of 0°) in the transmitting and receivingarray antenna arrangements (with De=0.5λ, Dt=1.5λ, and Dr=λ) illustratedin FIG. 6. As illustrated in FIG. 7, no grating lobe is generated in anangle range of ±90° of the main beam direction.

As described above, in the present embodiment, the transmitting antennas106 and the receiving antennas 202 are arranged such that the difference(absolute value) between the element spacing of the transmitting arrayantenna including the transmitting antennas 106 and the element spacingof the receiving array antenna including the receiving antennas 202 isequal to the predetermined element spacing with which no grating lobe isgenerated.

In this manner, the element spacing of the virtual receiving arrayconfigured in accordance with the arrangement relation between thetransmitting antennas 106 and the receiving antennas 202 can be set tothe predetermined element spacing with which no grating lobe isgenerated. This can prevent generation of a false detection due to anygrating lobe when the direction estimation processing is performed atthe direction estimator 214.

Thus, the present embodiment can prevent generation of an unnecessarygrating lobe, thereby achieving a desired directivity pattern, even whenarray elements having a subarray configuration are arranged.

FIG. 6 illustrates an exemplary configuration in which each arrayantenna has a linear arrangement in the horizontal direction so as toperform the arrival direction estimation in the horizontal direction.However, in the present embodiment, even when the array antenna has alinear arrangement in the vertical direction, the virtual receivingarray having the predetermined element spacing with which no gratinglobe is generated can be arranged in the vertical direction as well soas to perform the arrival direction estimation in the verticaldirection.

[Variation 1]

Variation 1 describes an example in which the arrival directionestimation is performed in both of the horizontal and verticaldirections.

Transmitting array elements and receiving array elements aretwo-dimensionally arranged in the vertical and horizontal directions.

FIG. 8 illustrates the antenna arrangement of a transmitting arrayincluding Nt=6 transmitting antennas 106 (Tx#1 to Tx#6), the antennaarrangement of a receiving array including Na=3 receiving antennas 202(Rx#1, Rx#2, and Rx#3), and the antenna arrangement of a virtualreceiving array (including Nt×Na=18 elements) configured in accordancewith these transmitting and receiving array antennas.

In FIG. 8, the transmitting array has a two-dimensional arrangement oftwo subarray elements in the horizontal direction and three subarrayelements in the vertical direction.

In FIG. 8, the horizontal dimension of each subarray element is equal toD_(subarray), and the vertical dimension is equal to or smaller than De.In other words, the horizontal dimension of each antenna element islarger than the predetermined antenna element spacing De and thevertical dimension is equal to or smaller than the predetermined antennaelement spacing De.

FIG. 8 illustrates an example in which the predetermined antenna elementspacing De is λ/2, the subarray element spacing Dt of the transmittingarray antenna in the horizontal direction is 1.5λ, and the elementspacing of the transmitting array antenna in the vertical direction isDe. The subarray element spacing Dr of the receiving array antenna inthe horizontal direction is λ.

In this case, as illustrated in FIG. 8, the absolute value of thedifference between the subarray element spacing Dt of the transmittingarray antenna and the subarray element spacing Dr of the receiving arrayantenna is equal to the predetermined antenna element spacing De in thehorizontal direction. As illustrated in FIG. 8, the element spacing ofthe transmitting array antenna is equal to the predetermined antennaelement spacing De in the vertical direction.

Accordingly, as illustrated in FIG. 8, the element spacing of thecentral part (other than edges) of the virtual receiving array is equalto the predetermined antenna element spacing De (=|Dt−Dr|=λ/2) in thehorizontal direction.

As illustrated in FIG. 8, the element spacing of the virtual receivingarray is equal to the predetermined antenna element spacing De in thevertical direction like the element spacing of the transmitting array inthe vertical direction.

In other words, the virtual receiving array has an array arrangementwith which no grating lobe is generated in the radar detection anglerange both in the horizontal and vertical directions.

As expressed in an expression below, for the arrival directionestimation in the horizontal and vertical directions, the directionestimator 214 calculates the direction estimation evaluating functionvalue P(θ_(u), φ_(v), k, fs, w) by treating the azimuth direction θ_(u)and the elevation direction φ_(v) as variables, and sets the azimuth andelevation directions at which the maximum value of the directionestimation evaluating function value is obtained, as an arrivaldirection estimation value DOA(k,fs,w).

$\begin{matrix}{{D\; O\; {A\left( {k,{fs},w} \right)}} = {\arg \; {\max\limits_{\theta_{u},\varphi_{v}}{P\left( {\theta_{u},\varphi_{v},k,{fs},w} \right)}}}} & (11)\end{matrix}$

In this expression, u=1, . . . , NU. The arg max P(x) is an operatorthat outputs a domain value at which the function value P(x) is atmaximum.

Different kinds of the evaluating function value P(θ_(u), φ_(v), k, fs,w) are provided by different arrival direction estimation algorithms.For example, an estimation method using an array antenna disclosed inCadzow described above may be used. For example, the beamformer methodcan be expressed in an expression below. Alternatively, the Capon andMUSIC methods are applicable as well.

P(θ_(u), φ_(v), k, fs,w)=a(θ_(u), φ_(v))^(H)H_(—after—cal)(k,fs,w)a(θ_(u), φ_(v))   (12)

In this expression, a superscript H is the Hermite transpositionoperator, and a(θ_(u), φ_(v)) represents the directional vector forarrival wave in the azimuth direction θ_(u) and the elevation directionφ_(v).

FIGS. 9A and 9B illustrate directivity patterns (Fourier beam patternswith the main beam at the direction of)0° of the transmitting andreceiving array antenna arrangements (with De=0.5λ, Dt=1.5λ, and Dr=1λ)illustrated in FIG. 8 in the horizontal and vertical directions,respectively.

As illustrated in FIG. 9A, no grating lobe is generated in the anglerange of ±90° of the main beam direction in the horizontal direction. Inaddition, as illustrated in FIG. 9B, a beam pattern in which no gratinglobe is generated is formed in the vertical direction.

Use of such arrangements of the transmitting and receiving arrayantennas can prevent generation of a false detection due to any gratinglobe in both of the horizontal and vertical directions when thedirection estimation processing is performed at the direction estimator214.

Thus, Variation 1 can prevent generation of an unnecessary grating lobe,thereby achieving a desired directivity pattern, even when arrayelements having a subarray configuration are two-dimensionally arranged.

Although FIG. 8 illustrates the case in which the dimension of thesubarray element in the horizontal direction is D_(subarray) (>De),Variation 1 is also applicable when the dimension of the subarrayelement in the vertical direction is D_(subarray) (>De). In this case,the transmitting array may be arranged in the vertical direction suchthat the difference (absolute value) between the element spacing of thetransmitting array antenna and the element spacing of the receivingarray antenna is equal to the predetermined element spacing with whichno grating lobe is generated.

[Variation 2]

Variation 2 describes another example in which the arrival directionestimation is performed in both of the horizontal and verticaldirections.

Specifically, when the transmitting array antenna has an element spacingequal to Dt (>De) in the horizontal direction and an element spacingequal to the predetermined antenna element spacing De in the verticaldirection, in the transmitting array antenna, arrays adjacent to eachother in the vertical direction, each array having two subarray elementslinearly arranged in the horizontal direction, are shifted from eachother in the horizontal direction by the predetermined antenna elementspacing De.

FIG. 10 illustrates the antenna arrangement of a transmitting arrayincluding Nt=6 transmitting antennas 106 (Tx#1 to Tx#6), the antennaarrangement of a receiving array including Na=3 receiving antennas 202(Rx#1, Rx#2, and Rx#3), and the antenna arrangement of a virtualreceiving array (including Nt×Na=18 elements) configured in accordancewith these transmitting and receiving array antennas.

In FIG. 10, the transmitting array has a two-dimensional arrangement oftwo subarray elements in the horizontal direction and three subarrayelements in the vertical direction.

In FIG. 10, the dimension of the subarray element in the horizontaldirection is equal to D_(subarray), and the dimension of the subarrayelement in the vertical direction is equal to or smaller than De. Inother words, the dimension of each antenna element is larger than thepredetermined antenna element spacing De in the horizontal direction andequal to or smaller than the predetermined antenna element spacing De inthe vertical direction.

In FIG. 10, similarly to FIG. 8, the predetermined antenna elementspacing De is λ/2, the subarray element spacing Dt of the transmittingarray antenna in the horizontal direction is 1.5λ, and the elementspacing of the transmitting array antenna in the vertical direction isDe. The subarray element spacing Dr of the receiving array antenna inthe horizontal direction is λ.

Similarly to Variation 1 (FIG. 8), as illustrated in FIG. 10, theabsolute value of the difference between the subarray element spacing Dtof the transmitting array antenna and the subarray element spacing Dr ofthe receiving array antenna is equal to the predetermined antennaelement spacing De in the horizontal direction. As illustrated in FIG.10, the element spacing of the transmitting array antenna is equal tothe predetermined antenna element spacing De in the vertical direction.

In FIG. 10, the transmitting antennas 106 disposed away from each otherby the antenna element spacing De in the vertical direction of thetransmitting array antenna (the transmitting antennas 106 adjacent toeach other in the vertical direction) are shifted from each other in thehorizontal direction by the antenna element spacing De. In other words,in the transmitting array antenna, the arrays adjacent to each other inthe vertical direction, each array having two subarray elements linearlyarranged in the horizontal direction, are shifted from each other in thehorizontal direction by the predetermined element spacing.

For example, in FIG. 10, the array (in other words, a subarray elementarray) of the transmitting antennas Tx#1 and Tx#2 and the array of thetransmitting antennas Tx#3 and Tx#4 adjacent to the array of thetransmitting antennas Tx#1 and Tx#2 in the vertical direction areshifted from each other by the antenna element spacing De. Similarly,the array of the transmitting antennas Tx#3 and Tx#4 and the array ofthe transmitting antennas Tx#5 and Tx#6 adjacent to the array of thetransmitting antennas Tx#3 and Tx#4 in the vertical direction areshifted from each other in the horizontal direction by the antennaelement spacing De.

In FIG. 10, the element spacing of the central part (other than edges)of the virtual receiving array is equal to the predetermined antennaelement spacing De (=|Dt−Dr|=λ/2) in the horizontal direction. Asillustrated in FIG. 10, similarly to the element spacing of thetransmitting array in the vertical direction, the element spacing of thevirtual receiving array is equal to the predetermined antenna elementspacing De in the vertical direction. In other words, the virtualreceiving array has an array arrangement with which no grating lobe isgenerated in the radar detection angle range.

As illustrated in FIG. 10, the array of array elements in the centralpart (second row) of the virtual receiving array in the verticaldirection is shifted from the other arrays of array elements (first andthird rows) by De in the horizontal direction. Accordingly, in FIG. 10,the antenna elements are arranged on a two-dimensional plane on whichthe virtual receiving array is arranged, at a narrower spacing than thatof Variation 1 (FIG. 8). This allows the virtual receiving array toachieve a reduced sidelobe level.

FIGS. 11A and 11B illustrate directivity patterns (Fourier beam patternswith the main beam at the direction of)0° of the transmitting andreceiving array antenna arrangements (with De=0.5λ, Dt=1.5λ, and Dr=λ)illustrated in FIG. 10 in the horizontal and vertical directions,respectively.

As illustrated in FIG. 11A, no grating lobe is generated in the anglerange of ±90° of the main beam direction in the horizontal direction. Inaddition, as illustrated in FIG. 11B, a beam pattern in which no gratinglobe is generated is formed in the vertical direction.

Moreover, as illustrated in FIG. 11A, the directivity pattern in thehorizontal direction has reduced sidelobe levels as compared to those ofVariation 1 (FIG. 9A).

Use of such arrangements of the transmitting and receiving arrayantennas can prevent generation of a false detection due to a gratinglobe and a sidelobe in both of the horizontal and vertical directionswhen the direction estimation processing is performed at the directionestimator 214.

Thus, Variation 2 can prevent generation of an unnecessary grating lobeand achieve a reduced sidelobe level, thereby achieving a desireddirectivity pattern, even when array elements having a subarrayconfiguration are two-dimensionally arranged.

[Variation 3]

Variation 3 describes another example in which the arrival directionestimation is performed in both of the horizontal and verticaldirections.

Specifically, in the transmitting array antenna, arrays adjacent to eachother in the vertical direction, each array having subarray elementslinearly arranged in the horizontal direction, are arranged in thevertical direction at a spacing equal to the product of thepredetermined antenna element spacing De and a constant α, and thearrays adjacent to each other in the vertical direction, each arrayhaving two subarray elements linearly arranged in the horizontaldirection, are shifted from each other in the horizontal direction by aspacing equal to the product of the predetermined antenna elementspacing De and a constant β.

FIG. 12 illustrates the antenna arrangement of a transmitting arrayincluding Nt=6 transmitting antennas 106 (Tx#1 to Tx#6), the antennaarrangement of a receiving array including Na=3 receiving antennas 202(Rx#1, Rx#2, and Rx#3), and the antenna arrangement of a virtualreceiving array (including Nt×Na=18 elements) configured in accordancewith these transmitting and receiving array antennas.

In FIG. 12, the transmitting array has a two-dimensional arrangement oftwo subarray elements in the horizontal direction and three subarrayelements in the vertical direction.

In FIG. 12, the dimension of the subarray element in the horizontaldirection is equal to D_(subarray), and the dimension of the subarrayelement in the vertical direction is equal to or smaller than De. Inother words, the dimension of each antenna element is larger than thepredetermined antenna element spacing De in the horizontal direction andequal to or smaller than the predetermined antenna element spacing De inthe vertical direction.

In FIG. 12, similarly to FIG. 8, the predetermined antenna elementspacing De is λ/2, the subarray element spacing Dt of the transmittingarray antenna in the horizontal direction is 1.5λ, and the subarrayelement spacing Dr of the receiving array antenna in the horizontaldirection is λ. The subarray element spacing Dr of the receiving arrayantenna in the horizontal direction is λ.

Similarly to Variations 1 and 2 (FIGS. 8 and 10), as illustrated in FIG.12, the absolute value of the difference between the subarray elementspacing Dt of the transmitting array antenna and the subarray elementspacing Dr of the receiving array antenna is equal to the predeterminedantenna element spacing De in the horizontal direction.

As illustrated in FIG. 12, the element spacing of the transmitting arrayantenna in the vertical direction is a spacing αDe as the product of thepredetermined antenna element spacing De and the constant α.

In FIG. 12, the transmitting antennas 106 disposed away from each otherby the element spacing αDe in the vertical direction of the transmittingarray antenna (the transmitting antennas 106 adjacent to each other inthe vertical direction) are shifted from each other in the horizontaldirection by a spacing βDe as the product of the predetermined antennaelement spacing De and the constant β. In other words, in thetransmitting array antenna, the arrays adjacent to each other in thevertical direction, each array having the two subarray elements linearlyarranged in the horizontal direction, are shifted from each other in thehorizontal direction by a spacing β times larger than the predeterminedelement spacing.

For example, in FIG. 12, the array of the transmitting antennas Tx#1 andTx#2 and the array of the transmitting antennas Tx#3 and Tx#4 adjacentto the array of the transmitting antennas Tx#1 and Tx#2 in the verticaldirection are shifted from each other by the spacing βDe. Similarly, thearray of the transmitting antennas Tx#3 and Tx#4 and the array of thetransmitting antennas Tx#5 and Tx#6 adjacent to the array of thetransmitting antennas Tx#3 and Tx#4 in the vertical direction areshifted from each other in the horizontal direction by the spacing μDe.

For example, α=(3)^(0.5)/2≈0.866, and β=0.5.

In FIG. 12, the element spacing of the central part (other than edges)of the virtual receiving array is equal to the predetermined antennaelement spacing De (=|Dt−Dr|=λ/2) in the horizontal direction.

As illustrated in FIG. 12, similarly to the element spacing of thetransmitting array in the vertical direction, the element spacing of thevirtual receiving array in the vertical direction is αDe (=(3)^(0.5)De).

In other words, the virtual receiving array has an array arrangementwith which no grating lobe is generated in the radar detection anglerange.

As illustrated in FIG. 12, the array of array elements in the centralpart (second row) of the virtual receiving array in the verticaldirection is shifted from the other arrays of array elements (first andthird rows) by βDe (=0.5De) in the horizontal direction.

Accordingly, in FIG. 12, similarly to Variation 2 (FIG. 10), antennaelements are arranged on the two-dimensional plane on which the virtualreceiving array is arranged, at a narrower spacing than that ofVariation 1 (FIG. 8). This allows the virtual receiving array to achievea reduced sidelobe level.

As illustrated in FIG. 12, each distance between three antenna elementsin the central part of the virtual receiving array adjacent to eachother on the two-dimensional plane on which the virtual receiving arrayis arranged is the predetermined antenna element spacing De. In otherwords, straight lines connecting three array elements adjacent to eachother on the two-dimensional plane on which the virtual receiving arrayis arranged form an equilateral triangle having all three sides equal tothe antenna element spacing De. This equilateral triangle latticearrangement provides a high grating lobe preventing performance ascompared to that provided by a rectangular lattice arrangement for thesame opening length. Thus, Variation 3 can further prevent generation ofa grating lobe and achieve a further reduced sidelobe level as comparedto Variation 2.

To do so, the constants α and β need to be set such that three arrayelements adjacent to each other in two dimensions of the vertical andhorizontal directions are arranged at the predetermined antenna elementspacing De (in an equilateral triangle having all three sides equal toDe).

FIGS. 13A and 13B illustrate directivity patterns (Fourier beam patternswith the main beam at the direction of 0°) of the transmitting andreceiving array antenna arrangements (with De=0.5λ, Dt=1.5λ, Dr=1λ,α=(3)^(0.5)/2, and β=0.5) illustrated in FIG. 12 in the horizontal andvertical directions, respectively.

As illustrated in FIG. 13A, no grating lobe is generated in the anglerange of ±90° of the main beam direction in the horizontal direction. Inaddition, as illustrated in FIG. 13B, a beam pattern in which no gratinglobe is generated is formed in the vertical direction, too.

Moreover, as illustrated in FIG. 13A, the directivity pattern in thehorizontal direction has reduced sidelobe levels as compared to those ofVariation 1 (FIG. 9A).

FIG. 13A also shows that sidelobe levels appearing in the directionclosest to the main lobe (the direction of ±30° in FIG. 13A) in thedirectivity pattern of the horizontal direction are reduced incomparison with Variation 2 (FIG. 11A).

Use of such arrangements of the transmitting and receiving arrayantennas can prevent generation of a false detection due to a gratinglobe and a sidelobe in both of the horizontal and vertical directionswhen the direction estimation processing is performed at the directionestimator 214.

Thus, Variation 3 can prevent generation of an unnecessary grating lobeand achieve a reduced sidelobe level even when array elements having asubarray configuration are two-dimensionally arranged, thereby achievinga desired directivity pattern.

The above has described the embodiment according to an aspect of thepresent disclosure.

The operations according to the embodiments and variations may beperformed in combination as appropriate.

The embodiment above describes the example in which the number Nt of thetransmitting antennas 106 is two or three, and the number Na of thereceiving antennas 202 is three. However, the number Nt of thetransmitting antennas 106 and the number Na of the receiving antennas202 are not limited to these numbers.

Although the embodiment above describes the case in which thetransmitting antennas 106 and the receiving antennas 202 are each asubarray element including two antenna elements, the transmittingantennas 106 and the receiving antennas 202 may each include threeantenna elements or more.

Variations 1 to 3 of the embodiment above each describe the case inwhich the transmitting array antenna has a two-dimensional arrangementin the horizontal and vertical directions, and the receiving arrayantenna has a one-dimensional arrangement in the horizontal direction.However, in the present disclosure, the receiving array antenna may havea two-dimensional arrangement, and the transmitting array antenna mayhave a one-dimensional arrangement. In this case, the above-describedarrangement of subarray elements in the transmitting array antenna maybe applied to the arrangement of subarray elements in the receivingarray antenna.

The embodiment above describe the case in which the dimension of eachantenna element is larger than the predetermined antenna element spacingDe in the horizontal direction and equal to or smaller than thepredetermined antenna element spacing De in the vertical direction.However, the vertical dimension of the antenna element may be largerthan the predetermined antenna element spacing De and the horizontaldimension may be equal to or smaller than the predetermined antennaelement spacing De. In this case, the above-described arrangements ofsubarray elements in the transmitting and receiving array antennas maybe exchanged between the horizontal direction and the verticaldirection.

Although the embodiments describes a case of using a coded pulse radar,the present disclosure is also applicable to a radar type such as achirp pulse radar using frequency-modulated pulsed waves.

In the radar device 10 illustrated in FIG. 2, the radar transmitter 100and the radar receiver 200 may be individually arranged at physicallyseparated locations.

The above description is made on the radar device having a configurationin which the radar transmitter sends out different transmission signalsprovided with code division multiplexing through multiple transmittingantennas, and then the radar receiver separates the transmission signalsto perform reception processing thereon. However, the configuration ofthe radar device is not limited thereto, and may be such that the radartransmitter sends out different transmission signals provided withfrequency division multiplexing through multiple transmitting antennas,and then the radar receiver separates the transmission signals toperform reception processing thereon. Alternatively, the radar devicemay have a configuration in which the radar transmitter sends outdifferent transmission signals provided with time division multiplexingthrough multiple transmitting antennas, and then the radar receiverperforms reception processing on the transmission signals. Theseconfigurations provide the same effect as that of the embodiment above.

The radar device 10 includes, although not illustrated, a centralprocessing unit (CPU), a storage medium such as a read only memory (ROM)storing a control program, and, and a working memory such as a randomaccess memory (RAM). With this configuration, the function of eachcomponent described above is achieved through execution of the controlprogram by the CPU. The hardware configuration of the radar device 10is, however, not limited thereto. For example, each functional componentof the radar device 10 may be achieved as an integrated circuit (IC).The functional components may be provided as individual chips, or partor all thereof may be included in a single chip.

<Summary of the Present Disclosure>

A radar device according to the present disclosure includes: radartransmission circuitry which, in operation, transmits a radar signalthrough a transmitting array antenna at a predetermined transmissionperiod; and radar reception circuitry which, in operation, receives areflected wave signal which is the radar signal reflected by an objectthrough a receiving array antenna. The transmitting array antenna andthe receiving array antenna each include multiple subarray elements. Thesubarray elements are linearly arranged in a first direction in each ofthe transmitting array antenna and the receiving array antenna. Eachsubarray element includes multiple antenna elements. A dimension of eachsubarray element in the first direction is larger than a predeterminedantenna element spacing. An absolute value of a difference between asubarray element spacing of the transmitting array antenna and asubarray element spacing of the receiving array antenna is equal to thepredetermined antenna element spacing.

In the radar device according to the present disclosure, thepredetermined antenna element spacing is equal to or larger than 0.5wavelength and equal to or smaller than 0.75 wavelength.

In the radar device according to the present disclosure, the subarrayelements in one of the transmitting array antenna and the receivingarray antenna are further arranged in a second direction orthogonal tothe first direction. The dimension of each subarray element in the firstdirection is larger than the predetermined antenna element spacing, theabsolute value of the difference between the subarray element spacing ofthe transmitting array antenna in the first direction and the subarrayelement spacing of the receiving array antenna in the first direction isequal to the predetermined antenna element spacing, the dimension of thesubarray element in the second direction is not larger than thepredetermined antenna element spacing, and the subarray elements arearranged in the second direction at the predetermined antenna elementspacing.

In the radar device according to the present disclosure, the subarrayelements arranged in the second direction are shifted from each other inthe first direction by the predetermined antenna element spacing.

In the radar device according to the present disclosure, the subarrayelements in one of the transmitting array antenna and the receivingarray antenna are further arranged in the second direction orthogonal tothe first direction. The dimension of each subarray element in the firstdirection is larger than the predetermined antenna element spacing, theabsolute value of the difference between the subarray element spacing ofthe transmitting array antenna in the first direction and the subarrayelement spacing of the receiving array antenna in the first direction isequal to the predetermined antenna element spacing in the firstdirection, the subarray elements are arranged in the second direction ata spacing ((√3)/2) times larger than the predetermined antenna elementspacing, and the subarray elements adjacent to each other in the seconddirection are shifted from each other in the first direction by aspacing (½) times larger than the predetermined antenna element spacing.

While the various embodiments (variations) have been described abovewith reference to the drawings, it is to be understood that the presentdisclosure is not limited to the disclosed exemplary embodiments. It isclear that the skilled person in the art could think of variousmodifications or corrections within the scope of the claims, which areunderstood to belong to the technical range of the present disclosure.The components described in the embodiments (variations) may beoptionally combined without departing from the spirit of the disclosure.

The present disclosure has an exemplary configuration using hardware inthe embodiments described above, but is achievable with software incombination with hardware.

Each functional block used in the description of the embodiments istypically achieved as an LSI, which is an integrated circuit. Theintegrated circuit may control the functional block used in thedescription of the embodiments, and include an input and an output.These integrated circuits may be provided as individual chips, or partor all thereof may be included in a single chip. The LSI may be calledan IC, system LSI, super LSI, or ultra LSI depending on its density ofintegration.

A method of the integration is not limited to LSI, but may be achievedusing a dedicated circuit or a general-purpose processor. Theintegration may be achieved using a field programmable gate array(FPGA), which is programmable after LSI manufacturing, or areconfigurable processor, in which connection or setting of circuitcells inside the LSI is reconfigurable.

Moreover, any novel integration technology provided to replace LSI bythe progress of the semiconductor technology or any technology derivingtherefrom may be used for integration of a functional block. A possiblereplacement is, for example, an application of biotechnology.

The present disclosure provides a preferable radar device that performsdetection in a wide-angle range.

What is claimed is:
 1. A radar device comprising: radar transmissioncircuitry which, in operation, transmits a radar signal through atransmitting array antenna at a predetermined transmission period; andradar reception circuitry which, in operation, receives a reflected wavesignal which is the radar signal reflected by an object through areceiving array antenna, wherein the transmitting array antenna and thereceiving array antenna each include a plurality of subarray elements,the subarray elements are linearly arranged in a first direction in eachof the transmitting array antenna and the receiving array antenna, eachsubarray element includes a plurality of antenna elements, a dimensionof each subarray element in the first direction is larger than apredetermined antenna element spacing, and an absolute value of adifference between a subarray element spacing of the transmitting arrayantenna and a subarray element spacing of the receiving array antenna isequal to the predetermined antenna element spacing.
 2. The radar deviceaccording to claim 1, wherein the predetermined antenna element spacingis not smaller than 0.5 wavelength and not larger than 0.75 wavelength.3. The radar device according to claim 1, wherein: the subarray elementsin one of the transmitting array antenna and the receiving array antennaare further arranged in a second direction orthogonal to the firstdirection, the dimension of each subarray element in the first directionis larger than the predetermined antenna element spacing, the absolutevalue of the difference between the subarray element spacing of thetransmitting array antenna in the first direction and the subarrayelement spacing of the receiving array antenna in the first direction isequal to the predetermined antenna element spacing, the dimension of thesubarray element in the second direction is not larger than thepredetermined antenna element spacing, and the subarray elements arearranged in the second direction at the predetermined antenna elementspacing.
 4. The radar device according to claim 3, wherein the subarrayelements arranged in the second direction are shifted from each other inthe first direction by the predetermined antenna element spacing.
 5. Theradar device according to claim 1, wherein: the subarray elements in oneof the transmitting array antenna and the receiving array antenna arefurther arranged in a second direction orthogonal to the firstdirection, the dimension of each subarray element in the first directionis larger than the predetermined antenna element spacing, the absolutevalue of the difference between the subarray element spacing of thetransmitting array antenna in the first direction and the subarrayelement spacing of the receiving array antenna in the first direction isequal to the predetermined antenna element spacing, the dimension of thesubarray element in the second direction is not larger than thepredetermined antenna element spacing, the subarray elements arearranged in the second direction at a spacing ((√3)/2) times larger thanthe predetermined antenna element spacing, and the subarray elementsadjacent to each other in the second direction are shifted from eachother in the first direction by a spacing (½) times larger than thepredetermined antenna element spacing.